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ADP3810AR-84

Secondary Side, Off-Line Battery Charger Controllers

Analog Devices 

ADP3810/ADP3811

+VIN

R1

80.6kΩ

VREF

VCTRL

VCTRL &

VREF RTN

R2

20kΩ

VIN

RTN

0.1µF

IRF7201

VBAT

1µF

10kΩ

0.1µF

0.1µF

VCC

VSENSE

OUT

VREF

ADP3811 VCS

VCTRL

GND

COMP

0.25Ω*

RC1

200Ω

CC1

1µF

RC2

560Ω

CC2

220nF

20kΩ*

2N3904

250Ω

220µF

1kΩ

R8

1kΩ

VBAT

=

2.0V

(

R––1 +

R2

1)

BATTERY

Figure 29. ADP3811 Controlling a Linear Battery Charger

The trade-off between using a linear regulator as shown versus

using a flyback or buck type of charger is efficiency versus sim-

plicity. The linear charger in Figure 29 is very simple, and it

uses a minimal amount of external components. However, the

efficiency is poor, especially when there is a large delta between

the input output voltages. The power loss in the pass transistor

is equal to (VIN–VBAT) × ICHARGE. Since the circuit is powered

from a wall adapter, efficiency may not be a big concern, but the

heat dissipated in the pass transistor could be excessive.

An important specification for this circuit is the dropout voltage,

which is the difference between the input and output voltage at

full charge current. There must be enough voltage to keep the

N-channel MOSFET on. In this case, the dropout voltage is

approximately 2.2 V for a 0.5 A output current. Two alternative

VIN

10kΩ

IRF7205

VBAT

ADP3811

VREF

2N3904

ADP3811

OUT

1kΩ

VIN

10kΩ

2N5058

2N3904

VBAT

1kΩ

ADP3811

OUT

2N3904

250Ω

a. P-Channel MOSFET

b. NPN Darlington

Figure 30. Alternative Pass Transistor for Linear Regulator

realizations of the pass element are shown in Figure 30. In case

(a), the pass transistor is a P-channel MOSFET. This provides

a lower dropout voltage so that VBAT can be within a few hun-

dred millivolts of VIN. In case (b), a Darlington configuration of

two npn transistors is used. The dropout voltage of this circuit

is approximately 2 V for a 0.5 A charge current.

STABILIZATION OF FEEDBACK LOOPS

The ADP3810/ADP3811 uses two transconductance error am-

plifiers with “merged” output stages to create a shared compen-

sation point (COMP) for both the current and voltage loops as

explained previously. Since the voltage and current loops have

significantly different natural crossover frequencies in a battery

charger application, the two loops need different inverted zero

feedback loop compensations that can be accomplished by two

series RC networks. One provides the needed low frequency

(typical fC < 100 Hz) compensation to the voltage loop, and the

other provides a separate high frequency (fC ~ 1 kHz–10 kHz)

compensation to the current loop. In addition, the current loop

input requires a ripple reduction filter on the VCS pin to filter

out switching noise. Instead of placing both RC networks on the

COMP pin, the current loop network is placed between VCS and

ground as shown in Figure 23 (CC2 and RC2). Thus, it performs

two functions, ripple reduction and loop compensation.

Loop Stability Criteria for Battery Charger Applications

1. The voltage loop has to be stable when the battery is

removed or floating.

2. The current loop has to be stable when the battery is being

charged within its specified charge current range.

3. Both loops have to be stable within the specified input source

voltage range.

Flyback Charger Compensation

Figure 31 shows a simplified form of a battery charger system

based on the off-line flyback converter presented in Figure 23.

With some modifications (no optocoupler, for example), this

model can also be used for converters such as a Buck Converter

(Figure 28) or a Linear Regulator (Figure 29). GM1 and GM2

are the internal GM amplifiers of the ADP3810/ADP3811, and

GM3 is the buffered output stage that drives the optocoupler.

The primary side in Figure 23 is represented here by the “Power

Stage,” which is modeled as GM4, a linear voltage controlled

current source model of the flyback transformer and switch.

The “Voltage Error Amplifier” block is the internal error ampli-

fier of the 3845 PWM-IC (RF = 3.3 kΩ in Figure 23), and it is

followed by an internal resistor divider. The optocoupler is

modeled as a current controlled current source as shown. Its

output current develops a voltage, VX, across RF. The gain val-

ues of all the blocks are defined below.

This linear model makes the calculation of compensation values

a manageable task. It also has the great benefit of allowing the

simulation of the ac response using a circuit simulator, such

as PSpice or MicroCap. For computer modeling, the GM

–12–

REV. 0

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